Nonlinear digital signal processor

ABSTRACT

A digital signal processor (DSP) comprises an input terminal configured to receive an input, an adaptive nonlinear phase filter coupled to the input terminal, the adaptive nonlinear phase filter having a time-varying phase response, and an adaptive nonlinear amplitude filter coupled to the input terminal, the adaptive nonlinear amplitude filter having a time-varying amplitude response. 
     A method of processing a signal comprises receiving the signal, sending the signal to an adaptive nonlinear phase filter, the adaptive nonlinear phase filter having a time-varying phase response, and sending the signal to an adaptive nonlinear amplitude filter, the adaptive nonlinear amplitude filter having a time-varying amplitude response.

CROSS REFERENCE TO OTHER APPLICATIONS

This application claims priority to U.S. Provisional Patent ApplicationNo. 60/848,089 entitled ADAPTIVE SELF-LINEARIZATION: LOW-POWER ANDLOW-COMPLEXITY SYSTEM OPERATION AND ARCHITECTURE, filed Sep. 29, 2006which is incorporated herein by reference for all purposes.

BACKGROUND OF THE INVENTION

Nonlinearity is a problem present in many signal processing systems. Forexample, the channel and the devices can introduce nonlinearity to atransmitted signal, thus causing distortion in the output. A typical wayof correcting the nonlinearity is by using a training signal with knownsignal characteristics such as amplitude, phase, frequency, datasequence, and modulation scheme. The nonlinearities in the system willintroduce distortion. The received signal is a composite signal of adistorted component, and an undistorted component that corresponds tothe ideal, undistorted training signal. During a training period, thetraining signal is available to the receiver. Filters in the receiver'ssignal processor are adjusted until the output matches the trainingsignal. This training technique requires that the ideal, undistortedtraining signal be available during the training period. The techniqueis sometimes impractical since adding the training to the manufacturingprocess will increase the cost of the device. Further, systemnonlinearities may vary due to factors such as variations in signalpaths, power supply, temperature, signal dynamics, Nyquist zone of thesignal, and/or aging of components. It is, however, often impractical tore-train the device since the undistorted training signal may no longerbe available. It would be desirable, therefore, to be able to moreeasily compensate for system nonlinearity. Some applications havegreater tolerance for the amount of time required to carry out thecompensation. Thus, it would also be useful to have low complexity andlow cost solutions for applications with less stringent timingrequirements.

BRIEF DESCRIPTION OF THE DRAWINGS

Various embodiments of the invention are disclosed in the followingdetailed description and the accompanying drawings.

FIG. 1A is a system diagram illustrating an embodiment of a system thatincludes an adaptive self-linearization module.

FIG. 1B is a system diagram illustrating an embodiment of a wirelessreceiver that includes an adaptive self-linearization module.

FIG. 2 is a flowchart illustrating an embodiment of a signal processingprocess.

FIGS. 3A-3C are frequency domain signal spectrum diagrams illustratingan example of nonlinear distortion in a signal.

FIG. 4A is a diagram illustrating an embodiment of an adaptiveself-linearization module.

FIG. 4B is a diagram illustrating an embodiment of a low latencyadaptive self-linearization system.

FIG. 5A is a flowchart depicting an embodiment of an adaptiveself-linearization process.

FIG. 5B is a flowchart illustrating another embodiment of an adaptiveself-linearization process.

FIG. 6 is a diagram illustrating details of an embodiment of an adaptivelinearization module.

FIG. 7 is a diagram illustrating an embodiment of a separation block.

FIG. 8 is a flowchart illustrating an embodiment of a process forextracting an undistorted component from a distorted signal.

FIG. 9 is a diagram illustrating the relative relationship of step sizeμ, number of taps N, and the type of linear component that can beeffectively extracted.

FIGS. 10A-10C are frequency domain signal diagrams illustrating anexample of a signal whose reference and target components occupydifferent frequency bands.

FIG. 11 is a block diagram illustrating another embodiment of anadaptive self-linearization module.

FIGS. 12A-12C are frequency domain signal diagrams illustrating anexample where both the reference component and the target componentoccupy multiple frequency bands.

FIG. 13 is a block diagram illustrating an embodiment of an adaptiveself-linearization system configured to correct a distorted signal (suchas 1230 of FIG. 12C) whose reference components and target componentsoccupy multiple separate frequency bands.

FIG. 14 is a signal diagram illustrating the operations of an exampleseparation block such as block 700 of FIG. 7.

FIG. 15A is a block diagram illustrating an embodiment of a simplifiedadaptive self-linearization module.

FIG. 15B is a block diagram illustrating another embodiment of asimplified adaptive linearization system.

FIG. 16 is a block diagram illustrating an example implementation of asimplified persistence filter embodiment.

FIG. 17 is a flowchart illustrating an embodiment of a process foradapting a simplified persistence filter.

FIG. 18 is a signal diagram illustrating the operations of a simplifiedseparation block employing a simplified persistence filter such asfilter 1600.

FIG. 19 is a block diagram illustrating an example implementation ofupdate block 1610 of FIG. 16.

FIG. 20 is a block diagram illustrating another simplified persistencefilter embodiment.

FIG. 21 is a block diagram illustrating an embodiment of a simplifiedadaptive DSP.

FIG. 22 is a block diagram illustrating an embodiment of a nonlinearadaptive phase filter.

DETAILED DESCRIPTION

The invention can be implemented in numerous ways, including as aprocess, an apparatus, a system, a composition of matter, a computerreadable medium such as a computer readable storage medium or a computernetwork wherein program instructions are sent over optical orcommunication links. In this specification, these implementations, orany other form that the invention may take, may be referred to astechniques. A component such as a processor or a memory described asbeing configured to perform a task includes both a general componentthat is temporarily configured to perform the task at a given time or aspecific component that is manufactured to perform the task. In general,the order of the steps of disclosed processes may be altered within thescope of the invention.

A detailed description of one or more embodiments of the invention isprovided below along with accompanying figures that illustrate theprinciples of the invention. The invention is described in connectionwith such embodiments, but the invention is not limited to anyembodiment. The scope of the invention is limited only by the claims andthe invention encompasses numerous alternatives, modifications andequivalents. Numerous specific details are set forth in the followingdescription in order to provide a thorough understanding of theinvention. These details are provided for the purpose of example and theinvention may be practiced according to the claims without some or allof these specific details. For the purpose of clarity, technicalmaterial that is known in the technical fields related to the inventionhas not been described in detail so that the invention is notunnecessarily obscured.

Signal linearization is described. As used herein, linearization refersto removing or compensating the nonlinearities in a signal. In someembodiments, based on an unknown distorted signal that is received,self-linearization is performed to compensate for nonlinear distortionand obtain an output signal that is substantially undistorted. As usedherein, self-linearization refers to calibration/linearization that doesnot require a training signal whose specific characteristics (such asfrequency components, amplitudes, phases, data sequence, and/ormodulation scheme) are already known to the module receiving the signal.In some embodiments, a digital signal processor comprising an adaptivenonlinear phase filter and an adaptive nonlinear amplitude filter isused.

FIG. 1A is a system diagram illustrating an embodiment of a system thatincludes an adaptive self-linearization module. An unknown input signalx is distorted by block 102, generating a distorted signal y. Block 102represents nonlinear distortion introduced by the transmission media,electronic circuits, or any other source. An adaptive self-linearizationmodule 102 is configured to correct for the distortion based on thereceived signal y.

FIG. 1B is a system diagram illustrating an embodiment of a wirelessreceiver that includes an adaptive self-linearization module. The systemis used to illustrate one application of the adaptive self-linearizationmodule, although many other applications and configurations exist. Inthe example shown, system 100 is a receiver. The system has a number ofcomponents including a radio frequency receiver, a filter, an amplifier,an analog to digital converter. Each of the components has somenonlinear characteristics, causing nonlinear distortion to the inputsignal. An adaptive self-linearization module 102 is configured tocorrect for nonlinearities in the receiver electronics, as well as thenonlinearities in the transmission channel. The adaptiveself-linearization module can also be used to correct nonlinearities inother systems where an input signal is distorted by nonlinearityintroduced by device components and/or transmission media. For example,the adaptive self-linearization module is sometimes included intransmitters, amplifiers, analog to digital converters, and many othertypes of electronic circuits to correct for system nonlinearities.

FIG. 2 is a flowchart illustrating an embodiment of a signal processingprocess. Process 200 may be implemented on adaptive self-linearizationmodule 102 of system 100. The process initiates when an unknown signalhaving an undistorted, ideal component and a distorted component isreceived (202). The signal is said to be unknown with respect to thereceiver of the signal since specific characteristics that define theundistorted component of the signal, such as amplitude, phase, signalfrequency, data sequence, or modulation scheme are not necessarilyavailable to the receiver. In other words, the receiver does notnecessarily have direct access to the undistorted component, nor is thereceiver necessarily able to reproduce the undistorted component withoutfurther linearization. Self-linearization, sometimes also referred to asblind linearization, is performed based on the received signal to obtainan output signal that is substantially similar to the undistortedcomponent (204). A training signal with known signal characteristics isnot required. Thus, the nonlinearities in the system can be correctedwhile the system is operating in the field. The linearization can bedone in real time since it requires no more than a few hundredmilliseconds from the time an unknown signal is received. The nonlinearcharacteristics of the system may change during operation due tononlinearity causing factors such as variations in the signal source,the paths, the power supply, temperature, signal dynamics, Nyquist zoneof the signal, sampling frequency, aging of components, component valuetolerances, etc. The adaptive self-linearization module can repeatedlyor continuously adapt to correct the nonlinearities despite changes inany of these factors. Further, the operation of the adaptiveself-linearization module is independent of the modulation scheme orencoding scheme of the received signal.

FIGS. 3A-3C are frequency domain signal spectrum diagrams illustratingan example of nonlinear distortion in a signal. In FIG. 3A, signal 300is an ideal, undistorted signal x centered at ω₀. Nonlinearcharacteristics of the system lead to distorted components, which areshown in FIG. 3B. The distorted components occur at integer multiples ofcenter frequency ω₀. The resulting signal to be received and processedby the adaptive self-linearization module is shown in FIG. 3C.

It is assumed that the distortion signal can be expressed using a Taylorseries. Even harmonics such as 304 and 306 are caused by distortionterms that are even powers of the signal (x², x⁴, etc.). The evenharmonics are relatively easy to remove since they are outside thefundamental frequency band of the desired signal. Odd harmonics such as303, 305, and 307 are caused by distortion terms that are odd powers ofthe signal (x³, x⁵, etc.). It is more difficult to remove the oddharmonics since harmonic 303 lies within the fundamental frequency bandof the desired signal. As will be shown in more detail below, theadaptive self-linearization module is able to approximately produce thedistorted components, thereby approximately determine the ideal,undistorted signal 300. Adaptive self-linearization can be performedbased on an unknown signal received while the device is operating (asopposed to using a known training signal). Further, an adaptiveself-linearization module allows the device to be calibrated regardlessof variations in the nonlinearity causing factors.

FIG. 4A is a diagram illustrating an embodiment of an adaptiveself-linearization module. In the example shown, module 400 includes anadaptive linearization module 402 and a delay component 404. Based onits input y_(n), the adaptive linearization module configures itsinternal filters to generate an output that approximates the distortedcomponent. Since the adaptation process leads to a delay of k samples inthe output, the output is denoted as η_(n−k). Details of how theadaptation is made are described below. y_(n) is sent to a delay moduleto obtain a delayed version, y_(n−k). Combiner 406 combines η_(n−k) fromy_(n−k) to obtain the desired, linearized signal component x_(n). Asused herein, combining may be addition or subtraction.

FIG. 5A is a flowchart depicting an embodiment of an adaptiveself-linearization process. Process 500 shown in the example may beimplemented on an adaptive self-linearization module such as 400. Duringthe process, an unknown distorted signal is separated into a referencecomponent and a target component (502). The reference component,sometimes referred to as the offending signal, includes an estimate ofone or more signal components that cause the nonlinear distortion in theunknown distorted signal. In some embodiments, the reference componentincludes an aggregated version of the undistorted component as well asthe harmonics within the frequency band of the undistorted component.The harmonics are relatively small and their effects can be ignored forpractical purposes. In some embodiments, the reference componentincludes one or more noise signals in a frequency band separate fromthat of the desired signal. The target component is the differencebetween the input signal and the reference component. A digital filteris adapted to generate a replica distortion signal that is substantiallysimilar to the distorted component. The adaptation is based at least inpart on the reference component and the target component (504). Byseparating the reference and target components, the system can train itsfilter based on a received signal whose characteristics are not knownprior to the training. The replica distortion signal is subtracted fromthe unknown distorted signal to generate the distortion corrected output(506).

FIG. 6 is a diagram illustrating details of an embodiment of an adaptivelinearization module. In the example shown, system 600 includes aseparation block 602 and an adaptive filter block 612. y_(n) is areceived signal with distortion. The signal is sent to separation block602, which includes a persistence filter 604 and a nonlinear signalextractor 605. As will be shown in more detail below, the separationblock is configured to extract from the input signal y_(n) a referencecomponent ŷ_(n). In this example, ŷ_(n) is a linearly enhanced versionof the input signal. The target component η is a function of thereceived signal and its history. At each time instance, η_(n) isexpressed as y_(n)−ŷ_(n).

For example, let the received signal y_(n)=1.001 x_(n)+0.01 x_(n) ³,where x_(n) is the desired undistorted component, and 0.001 x_(n)+0.01x_(n) ³ is the distorted component. A properly configured separationfilter will produce a reference component ŷ_(n) that is approximatelykx_(n) (k being a value close to 1), and a target component η_(n) thatis y_(n)−kx_(n).

In some embodiments, the nonlinear signal extractor further includes adelay element to give the input the same amount of delay as theseparation filter. In some embodiments, the nonlinear signal extractoroptionally includes a band pass filter, a low pass-filter, or a highpass filter. The additional filter is appropriate, for example, inapplications where the frequency band of the reference component isknown.

Returning to FIG. 6, ŷ_(n) and η_(n) are both sent to an adaptive filterblock 612, which includes an adaptive nonlinear digital signal processor(DSP) 608. The adaptive nonlinear DSP is sometimes implemented using anadaptive nonlinear filter. DSP 608 may be implemented using any suitabletechniques, such as techniques described in U.S. Pat. No. 6,856,191 byBatruni entitled “NONLINEAR FILTER” and U.S. Pat. No. 6,999,510 byBatruni entitled “NONLINEAR INVERSION”, which are herein incorporated byreference for all purposes. The patents incorporated by referencedescribe techniques for building nonlinear filters using linearelements, and for adapting such nonlinear filters to achieve desiredtransfer characteristics.

The DSP's inputs include the reference component ŷ_(n) and a feedbackerror signal e_(n) that is the difference between the target componentη_(n) and the DSP's output {circumflex over (η)}_(n). The DSP isconfigured to use ŷ_(n) as its input and η_(n) as its training signal toadapt its filter coefficients and drive the error signal to apredetermined level. The filter coefficients of the DSP's digitalfilters may be adapted using adaptive techniques including Least MeanSquares (LMS), Recursive Least Squares (RLS), or any other suitableadaptive techniques. The DSP adapts to implement a filter having atransfer function that is approximately the same as the nonlineartransfer function of the system, so that eventually the DSP's output{circumflex over (η)}_(n) is about the same as η_(n). In other words,the DSP's adapted transfer function approximately corresponds to thetransfer function representing the relationship of the distortedcomponent with respect to the undistorted component. Assuming that thedistorted component at the fundamental frequency is relatively small(e.g., 0.001 x_(n) as in the example discussed above), its effect isnegligible and therefore is for all practical purposes ignored. In theabove example, DSP 608 will adapt its filter parameters such that atransfer function of approximately 0.01 x_(n) ³ is obtained.

In the embodiment shown, the error signal of the DSP is expressed as:e _(n)=η_(n) −W _(n) ^(T) Ŷ _(n)  (1)where W_(n) ^(T)=[w_(n) w_(n−1) . . . w_(n−N+1) w_(n−N)] are thenonlinear coefficients andŶ_(n) ^(T)=[ŷ_(n) ŷ_(n−1) . . . ŷ_(n−N+1) ŷ_(n−N)] is the nonlinearfilter's input vector.

The nonlinear coefficients are expressed using the following generalform:

$\begin{matrix}\begin{matrix}{w_{n} = {{a_{n}{\hat{y}}_{n}} + b_{n} + {\sum\limits_{j = 1}^{K}{c_{j,n}{{{A_{j,n}^{T}{\hat{Y}}_{n}} + \beta_{j,n}}}}}}} \\{= {{a_{n}{\hat{y}}_{n}} + b_{n} + {\sum\limits_{j = 1}^{K}{{c_{j,n}\left( {{A_{j,n}^{T}{\hat{Y}}_{n}} + \beta_{j,n}} \right)}\lambda_{j,n}}}}}\end{matrix} & (2)\end{matrix}$whereλ_(j,n)=sign(A _(j,n) ^(T) Ŷ _(n)+β_(j,n))  (3)Ŷ_(n)[ŷ_(n+M)ŷ_(n+M−1) . . . ŷ_(n) . . . ŷ_(n−M+1)ŷ_(n−M])  (4)A_(j,n) ^(T)=[α_(M,n)α_(M−1,n) . . . α_(0,n) . . .α_(−M+1,n)α_(−M,n)]  (5)

The coefficients have a time index n because the filter is adaptive andtherefore time-varying. The nonlinear coefficients are adapted asfollows:A _(j,n+1) ^(T) =A _(j,n) ^(T) +μc _(j,n)λ_(j,n) Ŷ _(n) e _(n) ŷ_(n)  (6)β_(j,n+1)=β_(j,n) +μc _(j,n)+λ_(j,n) e _(n) ŷ _(n)  (7)c _(j,n+1) =c _(j,n) +μ|A _(j,n) ^(T) Ŷ _(n)+β_(j,n) |e _(n) ŷ _(n)  (8)α_(j,n+1)=α_(j,n) +μŷ _(n) e _(n) ŷ _(n)  (9)b _(j,n,+1) =b _(j,n) +μe _(n) ŷ _(n)  (10)

Returning to FIG. 6, separation block 602 employs persistence filter 604for separating the reference component from the received signal. Thepersistence filter is designed to boost the linear signal components andattenuate the noise and nonlinear signal components in the receivedsignal. An analogy to the persistence filter is a camera shutter, whichallows light to pass for a period of time in order to capture thestationary image. The background images that are non-stationary overthis period of time become blurry. Like a camera shutter, over a periodof time, the persistence filter captures the persistent portion of aninput signal and removes the non-persistent portion. The persistencefilter operates on pseudo stationary input signals that are not rapidlychanging (for example, a signal that is stationary for at least a fewmilliseconds). For a pseudo stationary input signal, the persistentportion is the average of the desired reference component, which isrelatively stable and enhances over time. In some embodiments, thepersistence filter is designed as an averaging, linear filter thatemphasizes the undistorted signal over noise, and emphasizes linearsignal components over nonlinear distortion.

FIG. 7 is a diagram illustrating an embodiment of a separation block. Inthis example, separation block 700 includes a persistence filter 702,which includes a delay line 704 to which the input y_(n) is sent, and aplurality of coefficient multipliers 706. The number of taps in thedelay line is represented as N=2K+1. In the example shown, K=512, whichmeans that the delay line has 1025 taps for delays of 0, 1, 2, . . .1024. Each y_(i) (i=n+512, n+511, . . . , n, . . . n−511, n−512) isscaled by multiplying with an adaptable coefficient v_(i). Themultiplication results are summed, producing the linear referencecomponent ŷ_(n). The center tap value y_(n) is selected, and ŷ_(n) issubtracted from y_(n) to produce an error ε_(n). In this case, ε_(n)corresponds to target η_(n). The error is fed back to updatecoefficients v_(i). An adaptive algorithm such as LMS or RLS is used toupdate the coefficients until ε_(n) approaches some predefined thresholdvalue. The separation block is configured to receive the input y_(n),and aggregate y_(n) over a period of time to produce an aggregate signalthat is substantially similar to the undistorted component. Theaggregate signal is considered substantially similar when ε_(n) meetssome predefined threshold value. The aggregate signal is then subtractedfrom the received input.

FIG. 8 is a flowchart illustrating an embodiment of a process forextracting an undistorted component from a distorted signal. Process 800may be implemented on a separation block, such as 700 shown in FIG. 7.In this example, during the process, a digital signal that includes anundistorted component and a distorted component is received (802). Aplurality of samples of the received signal are multiplied with aplurality of coefficients (804). The multiplication results are summedto produce an aggregate (805). The aggregate enhances the undistortedcomponent and attenuates the distorted component. An error is generatedby taking the difference between the aggregate and a sample of thereceived signal (806). The error is fed back to adapt the coefficients(808).

The persistence filter can be described using the following functions:η_(n) =y _(n) −V _(n) Y _(n)  (11)η_(n) =y _(n) −ŷ _(n)  (12)V _(n+1) =vV _(n)+μη_(n) Y  (13)where Y_(n)=[y_(n+K y) _(n+K−1) . . . y_(n) . . . y_(n−K−1) y_(n−K)], μis the adaptation step size that controls the persistency factor of thefilter and v is the forgetting factor that controls the speed with whichthe filter adapts to changing signal dynamics.

The number of filter taps N (also referred to as the order of thefilter) and the adaptive step size μ control the persistence filter'soperations. A given filter order and step size combination may beparticularly effective for emphasizing the received signal's linearcomponent within a certain range of bandwidth and amplitude. FIG. 9 is adiagram illustrating the relative relationship of step size μ, number oftaps N, and the type of linear component that can be effectivelyextracted. The diagram informs the choice of μ and N. Generally, ahigher N (i.e., a greater number of filter taps) should be used as theamplitude of the linear component goes down, and a smaller μ (i.e., asmaller step size) should be used as the bandwidth of the linearcomponent goes down. As shown in the diagram, if the linear componenthas a relatively large amplitude and a relatively narrow bandwidth (suchas signal 902), a persistence filter with a small μ and a small Nproduces good results. A linear component having a similarly largeamplitude but a wider bandwidth (signal 904) requires a relatively smallN and allows a greater μ. A small amplitude and large bandwidth linearcomponent (signal 906) requires a large N and a large μ. A smallamplitude and narrow bandwidth linear component (signal 908) requires asmall μ and a large N. During operation, N and μ can be adjusted to moreeffectively generate the emphasized linear component. For example, insome embodiments, a peak detector and a power level detector are used todetect the strength of the signal. The signal strength is a function ofthe signal's peak and bandwidth. Based on the detected signal strength,appropriate adjustments to N and μ are made according to systemrequirements to control the adaptation.

In some embodiments, the linearization process requires a large numberof samples. The delay k sometimes corresponds to hundreds or eventhousands of samples, resulting in delay on the order of tens or evenhundreds of milliseconds. Some applications (e.g. telecommunicationapplications) may require the linearization process to have a lowerlatency. FIG. 4B is a diagram illustrating an embodiment of a lowlatency adaptive self-linearization system. In the example shown, system420 is configured to have much lower latency than system 400. The DSPsshown in the system may be implemented as general or special purposeprocessors, or configurable filters. Adaptive linearization module 422configures an internal DSP to simulate the nonlinear transfer functionto be corrected and produces an output that is approximately equal tothe nonlinear residual signal. As discussed above, assuming that thedistortion within the fundamental frequency band is relatively small, asuccessfully adapted and configured DSP will have a transfer functionthat is approximately equal to the nonlinear transfer function to becorrected. The linearization module outputs the configurationparameters, w, to a shadow nonlinear DSP 424, which uses the parametersto configure its filters and duplicate the transfer function of the DSPemployed by the adaptive linearization module. DSP 424's latency L is onthe order of a few milliseconds, which is significantly smaller than thedelay due to adaptation k. As such, system 420 has significantly lessdelay than system 400.

FIG. 5B is a flowchart illustrating another embodiment of an adaptiveself-linearization process. Process 550 shown in the example may beimplemented on a low latency adaptive self-linearization module such as420. During the process, an unknown distorted signal is separated into areference signal and a target signal (552). A first digital filter isadapted to generate a replica distortion signal that is substantiallysimilar to the distorted component, where the adaptation is based atleast in part on the reference signal (554). A second digital filter isconfigured using coefficients from the adapted first digital filter(556). A second replica distortion signal that is substantially similarto the distorted component using the second digital filter (558).

In some embodiments, the reference component and the target componentoccupy separate frequency bands. FIGS. 10A-10C are frequency domainsignal diagrams illustrating an example of a signal whose reference andtarget components occupy different frequency bands. FIG. 10A shows theideal, undistorted component 1000, which is limited to frequency bandb₀. An example of the ideal signal is a radio frequency (RF) signal usedin a wireless communication system that employs some form of frequencydivision, where the signal occupies a specific frequency channel b₀.FIG. 10B shows the distortion component, which includes noise signalcomponent 1002 that is outside b₀, as well as harmonics of the noisecomponent, including 1004 which falls within frequency channel b₀, and1006 which lies outside b₀. An example of noise signal 1002 is anotherRF signal occupying an adjacent frequency channel relative to signal1000 and causing distortion in frequency channel b₀. FIG. 10C shows theresulting signal 1006. Although the general frequency ranges of thereference and target components are known, the specific characteristicsof the signal components are still unknown. Thus, the signal is suitablefor processing by any adaptive self-linearization module that implementsprocesses 200 or 500.

An adaptive self-linearization module such as 400 or 420 described abovecan be used to process the type of signal shown in FIG. 10C. Assumingthat the desired signal causes little distortion in its own frequencyband and that most of the distortion in the received signal is caused bynoise from neighboring frequency channel(s), it is possible to employadaptive self-linearization modules with less complex circuitry bytaking advantage of the fact that the reference and target componentsreside in different frequency bands. FIG. 11 is a block diagramillustrating another embodiment of an adaptive self-linearizationmodule. In the example shown, separation block 1102 includes a referencesignal band-specific filter 1104 and a target signal band-specificfilter 1114. In some embodiments, the reference band-specific filterincludes a band-stop filter configured to extract from the receivedsignal the noise component and its harmonics outside frequency band b₀and suppress the components within b₀, generating the referencecomponent ŷ_(n). The target signal band-specific filter includes aband-pass filter configured to pass components in frequency band b₀ andattenuate the rest of the frequencies, generating the target componentη_(n).

Based on reference component ŷ_(n), DSP adapts its parameters togenerate a replica of the distorted signal, {circumflex over (η)}_(n).The adaptation is possible because the reference component and thedistorted signal are correlated. {circumflex over (η)}_(n) is subtractedfrom the target component η_(n) to obtain the desired signal x_(n). Asuitable adaptation technique such as LMS or RLS is used to adapt theDSP. Some embodiments base the adaptation on equations (1)-(10).

Referring to FIGS. 10A-10C as an example, the input signal y_(n)corresponds to signal 1006. The separation block extracts referencecomponent ŷ_(n) which corresponds to components 1002 plus 1006 andtarget component η_(n) which corresponds to component 1008. In someembodiments, the separation block further limits the bandwidth ofreference component extraction such that only 1002 is extracted. Basedon ŷ_(n) and its feedback signal x_(n), the adaptive DSP adapts itstransfer function to generate {circumflex over (η)}_(n), whichapproximately corresponds to signal 1004

In some embodiments, the offending signals causing distortion in thefundamental frequency band of the desired signal may reside in multiplefrequency bands. FIGS. 12A-12C are frequency domain signal diagramsillustrating an example where both the reference component and thetarget component occupy multiple frequency bands. FIG. 12A shows theundistorted signal components 1200-1204, which occupy separate frequencybands b₁-b₃. FIG. 12B shows the distorted signal components, whichincludes several noise components 1210-1214 which reside outside b₁-b₃,and their harmonics 1216, 1218, and 1220 which reside within b₁, b₂, andb₃ respectively. FIG. 12C shows the resulting distorted signal 1230.

FIG. 13 is a block diagram illustrating an embodiment of an adaptiveself-linearization system configured to correct a distorted signal (suchas 1230 of FIG. 12C) whose reference components and target componentsoccupy multiple separate frequency bands. In the example shown, system1300 includes a reference component band-specific filter 1304 forselecting reference signal components ŷ_(n) that cause distortion (e.g.,signal components 1210-1214 shown in FIG. 12B). Filter 1304 may beimplemented using a plurality of bandpass filters. The system alsoincludes N target component band-specific filters for producing targetcomponents ηk_(n)(k=1, . . . , N) in specific frequency bands. In theexample shown in FIG. 12C, N=3, and target components corresponding to1232, 1234 and 1236 are produced. N DSPs are each adapted based on thereference component and a corresponding feedback signal xk_(n) togenerate distortion components {circumflex over (η)}k_(n)(k=1, . . . ,N). Each {circumflex over (η)}k_(n) is subtracted from the targetcomponent η_(n) to obtain the desired signal x_(n). The adaptationtechnique of each DSP is similar to what was described in FIG. 11.

Returning to the example shown in FIG. 6, where an adaptivelinearization module 600 is in some embodiments implemented usingseparation block 700 of FIG. 7. FIG. 14 is a signal diagram illustratingthe operations of an example separation block such as block 700 of FIG.7. In the diagram shown, a nonlinear signal 1400 is received andprocessed by the separation block. The length of the separation block'sdelay line is 1025. At time t=0, the first computation cycle begins.Samples y₀-y₁₀₂₄ are multiplied with corresponding adaptablecoefficients v₀-v₁₀₂₄, and the multiplication results are summed toproduce linear reference sample ŷ₅₁₂, which is subtracted from thecenter tap value y₅₁₂ to produce an error η₅₁₂ for updating theadaptable coefficients. At time t=1, the next computation cycle begins.Samples y₁-y₁₀₂₅ are multiplied with the updated coefficients v₀-v₁₀₂₄to produce linear reference sample ŷ₅₁₃, which is subtracted from thecenter tap value y₅₁₃ to produce an error η₅₁₃ that is used to updatethe adaptable coefficients. The separation block repeats the computationon every input clock cycle to generate a reference component sample andan error value, and continuously updates the adaptable coefficients.

The reference signal ŷ_(n) is sent to the adaptive filter block to trainthe DSP to generate a replica of the nonlinear component of the input.The DSP converges quickly because the separation block generates areference signal sample during each input clock cycle. Since the DSPwill continuously adapt to model system nonlinearities, the great amountof computation involved (especially the number of multiplications)demands high power consumption. Further, the implementation of theseparation block is more complex and efficient because of the number ofparallel multiplications. There are situations where the requirement ofconvergence speed is relaxed. For example, many electronic devices thatcould benefit from adaptive self-linearization only require theadaptation to be carried out during startup, and a few seconds of delayis tolerated. It would be useful if the separation block design could besimplified.

FIG. 15A is a block diagram illustrating an embodiment of a simplifiedadaptive self-linearization module. In the example shown, the principleof operation of module 1500 is similar to that of adaptive linearizationmodule 400 of FIG. 4A, where an unknown input signal y_(n) is received,and self-linearization is performed based at least in part on theunknown signal to obtain an output signal that is substantiallyundistorted. Separation block 1502, which includes a simplifiedpersistence filter 1504 and a nonlinear target component extractor 1506,separates the input signal into a linear reference component ŷ_(n) and anonlinear target component η_(n). η_(n) is fed back to the persistencefilter to adapt the filter coefficients. Adaptive filter block 1512updates a DSP 1508 based on the input signal and a feedback error signale_(n) that is generated based on the DSP output and the targetcomponent, to generate a replica of the nonlinear component of theinput, {circumflex over (η)}_(n).

The simplified adaptive linearization module operates at a lower rate tosave power. During adaptation, simplified persistence filter 1504receives the input signal y_(n) at full rate but computes one outputevery L samples, where L is an integer greater than 1. Thus, theadaptive linearization module operates at a rate of R/L. The output ofthe simplified persistence filter, reference signal ŷ_(n), is alsogenerated at a down sampled rate of R/L. For purposes of simplicity, inthe following discussion the input sampling rate R is assumed to be 1and the down sampled rate is therefore 1/L. The input to nonlineartarget component extractor 1506 is also down sampled at a rate of 1/L.In other words, one of every L input samples is sent to the extractor,which subtracts from its input the output of the simplified persistencefilter to generate the nonlinear target signal η_(n). DSP 1508 processesy_(n), and updates its filter coefficients at a rate of 1/L to generatethe nonlinear replica signal {circumflex over (η)}_(n). {circumflex over(η)}_(n) is compared with η_(n) using comparator 1510 to generate anerror signal e_(n), which is fed back to DSP to adapt its internalfilter coefficients. Since both the persistence filter and DSP operateat a slower rate, power consumption is reduced and chip circuitries areless complex.

FIG. 15B is a block diagram illustrating another embodiment of asimplified adaptive linearization system. In this example, the principleof operation of system 1550 is similar to that of adaptive linearizationsystem 420 of FIG. 4B. Module 1550 has a similar separation block 1552and adaptive filter block 1554 as module 1500. Separation block 1552includes a delay element 1556 that delays down-sampled input by L, and anonlinear target component extractor 1559 configured to subtract thereference signal ŷ_(n) from the delayed, down-sampled input to produce atarget signal η_(n). As discussed above, assuming that the distortionwithin the fundamental frequency band is relatively small, asuccessfully adapted and configured DSP will have a transfer functionthat is approximately equal to the nonlinear transfer function to becorrected. Once DSP A 1558 converges, the coefficients of the DSP engineare copied to shadow DSP B 1560. The input to DSP B is the distortedinput y_(n) without any delay, and the output of DSP B is a replicadistortion signal that is substantially similar to the distortedcomponent of the input signal. Thus, the replica distortion signalgenerated by DSP B is based at least in part on the target componentadaptively generated by DSP A. The replica distortion signal iscancelled directly from the input signal without delay. The resultingoutput signal of the system, x_(n), is a linearized signal with littledelay and low latency. Because components 1557, 1556, 1558 all operateat down-sampled rate 1/L, resources such as operators can be shared andpower consumption is reduced.

The simplified separation block is described using the followingequations:η_(n) =y _(n) −V _(n) Y _(n)  (14)η_(n) =y _(n) −ŷ _(n)  (15)V _(n+1) =vV _(n)+μη_(n) Y _(n)  (16)where input vector Y_(n)=[y_(n+K)y_(n+K−1) . . . y_(n) . . . y_(n−K−1)y_(n−K)] and filter coefficient vector V_(n)=[v_(n+K)v_(n+K−1) . . .v_(n) . . . v_(n−K−1)v_(n−K)]. In some embodiments, the initial value ofV_(n) is chosen such that only the center tap is 1 and the rest of thecoefficients are 0 (i.e., V_(n)=[0 0 . . . 1 . . . 0 0]). Other startingvalues are possible. μ is the adaptation step size that controls thepersistency factor of the filter and v is the forgetting factor thatcontrols the speed with which the filter adapts to changing signaldynamics. The values of μ and v are chosen depending on factors such assignal bandwidth, required speed of convergence, etc.

The next set of values is obtained L samples later. The simplifiedseparation block can be described using the following equations:η_(n+L) =y _(n+L) −V _(n+L) Y _(n+L)  (17)η_(n+L) =y _(n+L) −ŷ _(n+L)  (18)V _(n+L+1) =vV _(n+L)+μη_(n+L) Y _(n+L)  (19).

FIG. 16 is a block diagram illustrating an example implementation of asimplified persistence filter embodiment. In this example, filter 1600includes a memory 1602, implemented as a delay line, that is configuredto hold a total of 1025 input samples y_(n+512) to y_(n−512). The filteralso includes a memory 1604 that stores a total of 1025 adaptablecoefficients v_(n+512) to v_(n−512). The input samples and coefficientsare selected sequentially. In some embodiments, a switch or a similarselector component is used to make the selection on each clock cycle.For example, y_(n+512) and v_(n+512) are selected at time t=0, y_(n+511)and v_(n+511) at time t=2, y_(n+510) and v_(n+510) at time t=3, etc.Multiplier 1606 multiplies a selected pair of y_(n+i) and v_(n+i)values. Unlike the dedicated multipliers 706 shown in FIG. 7, multiplier1606 is referred to as a shared multiplier since it is shared by anumber of (y_(n+i), v_(n+i)) value pairs to carry out a plurality ofinput sample-coefficient multiplications sequentially over several clockcycles. The product is sent to an accumulator 1608, which accumulatesthe sum of the products over time, until all the (y_(n+i), v_(n+i))value pairs are computed and summed. In this example, 1025 samples willproduce a single reference signal sample ŷ_(n). ŷ_(n) is compared withthe value stored in the center tap of the delay line, y_(n). Theresulting error, which corresponds to the target nonlinear componentη_(n), is sent to an update block 1610 to update the coefficients.

FIG. 17 is a flowchart illustrating an embodiment of a process foradapting a simplified persistence filter. Process 1700 may beimplemented on a simplified persistence filter such as filter 1600 ofFIG. 16. At the beginning of the process, the filter is initialized atstep 1702. In this example, the initialization stage includes step 1704initializing the accumulator such that its sum S is 0, step 1706 loadingthe initial adaptable coefficients V_(n) into memory, and step 1708receiving and storing L input samples Y_(n) into memory. An inputsample-coefficient pair is selected at step 1710. The product of theselected input sample and the coefficient is computed by a sharedmultiplier at step 1712. The product is accumulated, in other wordsadded to the accumulator's sum at step 1714. Steps 1710-1714 repeatuntil all the input samples are processed, i.e., multiplied with acorresponding coefficient and their products accumulated. The sum of theaccumulator is the reference signal sample ŷ_(n). ŷ_(n) is compared withthe an input sample to generate the target nonlinear component η_(n) atstep 1716. In this example, the input sample is the center input sampley_(n). At step 1718, the next set of coefficients V_(n+1) are updatedbased on η_(n), V_(n), and Y_(n). In some embodiments, the update steptakes place while steps 1710-1714 are in progress. Once the coefficientsare updated, the process returns to step 1708. The accumulator is resetand a new set of L input samples are loaded into memory.

FIG. 18 is a signal diagram illustrating the operations of a simplifiedseparation block employing a simplified persistence filter such asfilter 1600. In the diagram shown, L=1025. At time=0, the firstprocessing cycle begins, where input samples y₀-y₁₀₂₄ are multipliedwith v₀-v₁₀₂₄. The products are accumulated and compared with the centertap y₅₁₂ to produce a linear reference sample ŷ₅₁₂ and nonlinear targetsignal η₅₁₂. At time=1025, the second processing cycle begins and inputsamples y₁₀₂₅-y₂₀₄₉ are sequentially multiplied with a set of updatedcoefficients. The products are accumulated and compared with the newcenter tap y₅₁₂ to produce another reference sample ŷ₁₅₃₆ and nonlineartarget signal η₁₅₃₆. Thus, the persistence filter produces outputsamples at a rate of 1/L of the input data rate. Although a lower datarate means that downstream processing components such as the DSP willconverge more slowly, using a shared multiplier to generate a lower datarate output reduces chip complexity and power consumption.

FIG. 19 is a block diagram illustrating an example implementation ofupdate block 1610 of FIG. 16. In this example, update block 1610 updatesthe coefficients v_(i) once for every L input samples. On each inputclock cycle, the selected (y_(n+i), v_(n+i)) pair and η_(n+L) (i.e.,η_(n) generated during the previous processing cycle, based on the lastL samples processed) are input to the update block. y_(n+i) ismultiplied with the adaptation step size μ and η_(n+L), and v_(n+i), ismultiplied with the forgetting factor v. The results are summed by anadder, and then delayed by L clock cycles to generate v_(n+L+i). Aswitch is used to update the storage location that currently stores thevalue v_(n+i), with the new v_(n+L+i) value.

FIG. 20 is a block diagram illustrating another simplified persistencefilter embodiment. In this example, persistence filter 2000 includes aplurality of shared operators. Each of the shared operators is used tooperate on a subset of the input samples and adaptable coefficients. Inthe example, the shared operators include shared multipliers 2002 eachconfigured to multiply k pairs of input samples with correspondingcoefficients over k clock cycles. Switches 2004 and 2006 are used toselect the appropriate input sample-coefficient pair on each clockcycle. The results are sent to accumulator 2010 to be summed. Areference signal sample ŷ_(n) and a nonlinear target signal sample{circumflex over (η)}_(n) are generated every k clock cycles. Each setof k coefficients are updated every k clock cycles via correspondingupdate blocks 2008. Update blocks 2008 may be implemented usingstructures similar to 1610 of FIG. 19, where k input samples and kcoefficients from the appropriate memory locations are selectively sentto each update structure.

In the example shown, k=4. The number of shared multipliers may vary fordifferent embodiments, and can be adjusted to increase or decrease theoutput rate, thereby fulfilling the timing requirements for convergencewhile minimizing power consumption and circuitry complexity.

Returning to FIGS. 15A and 15B, y_(n) and η_(n) are sent to an adaptivefilter block, which includes an adaptive nonlinear DSP configured toadaptively achieve a filter transfer function that approximates thesystem distortion transfer function describing the nonlinearcharacteristics of the channel. In systems 1500 and 1550, for example,the DSP's inputs include the reference component ŷ_(n) and a feedbackerror signal e_(n) that is the difference between the target componentη_(n) and the DSP's output {circumflex over (η)}_(n). The DSP isconfigured to use ŷ_(n) as its input and η_(n) as its training signal toadapt its filter coefficients and drive the error signal to apredetermined level.

The filter coefficients of the DSP's digital filters may be adaptedusing adaptive techniques including Least Mean Squares (LMS), RecursiveLeast Squares (RLS), or any other suitable adaptive techniques. The DSPadapts to implement a filter having a transfer function that isapproximately the same as the nonlinear transfer function of the system,so that eventually the DSP's output {circumflex over (η)}_(n) is aboutthe same as η_(n). In other words, the DSP's adapted transfer functionis approximately the same as the transfer function describing thenonlinear relationship of the distorted component with respect to theundistorted component.

The error signal of the DSP is generally expressed as:e _(n)=η_(n) −W _(n) ^(T) Ŷ _(n)=η_(n)−{circumflex over (η)}_(n)  (20)where {circumflex over (η)}_(n) is the nonlinear distortion replicasignal, W_(n) ^(T)=[w_(n)w_(n−1) . . . w_(n−N+1)w_(n−N)] are thenonlinear coefficients and Ŷ_(n) ^(T)=[ŷ_(n)ŷ_(n−1) . . .ŷ_(n−N+1)ŷ_(n−N)] is the nonlinear filter's input vector. In adaptivelinearization module embodiments such as 600 and 1500, where the DSP iscoupled to a separation block, the input vector of the DSP Ŷ_(n)corresponds to the linear reference component of the separation blockoutput.

The nonlinear coefficients of the DSP are expressed using the followinggeneral form:

$\begin{matrix}\begin{matrix}{w_{n} = {{a_{n}{\hat{y}}_{n}} + b_{n} + {\sum\limits_{j = 1}^{K}{c_{j,n}{{{A_{j,n}^{T}{\hat{Y}}_{n}} + \beta_{j,n}}}}}}} \\{= {{a_{n}{\hat{y}}_{n}} + b_{n} + {\sum\limits_{j = 1}^{K}{{c_{j,n}\left( {{A_{j,n}^{T}{\hat{Y}}_{n}} + \beta_{j,n}} \right)}\lambda_{j,n}}}}}\end{matrix} & (21)\end{matrix}$whereλ_(j,n)=sign(A _(j,n) ^(T) Ŷ _(n)+β_(j,n))  (22)Ŷ_(n)=[ŷ_(n=M)ŷ_(n+M−1) . . . ŷ_(n) . . . ŷ_(n−M−1)ŷ_(n−M])  (23)A_(j,n) ^(T)[α_(M,n)α_(M−1,n) . . . α_(0,n) . . .α_(−M+1,n)α_(−M,n)]  (24)

Coefficients α_(j,n), β_(j,n), c_(j,n), and λ_(j,n) have a time index nbecause the filter is adaptive and therefore time-varying. The startingvalues for the coefficients may be 0 or any small random number. Thenonlinear coefficients are adapted as follows:A _(j,n+1) ^(T) =A _(j,n) ^(T) +μc _(j,n)λ_(j,n) Ŷ _(n) e _(n) ŷ_(n)  (25)β_(j,n+1)=β_(j,n) +μc _(j,n)λ_(j,n) e _(n) ŷ _(n)  (26)c _(j,n+1) =c _(j,n) +μ|A _(j,n) ^(T) Ŷ _(n)+β_(j,n) |e _(n) ŷ_(n)  (27)a _(j,n+1) =a _(j,n) +μŷ _(n) e _(n) ŷ _(n)  (28)b _(j,n+1) =b _(j,n) +μe _(n) ŷ _(n)  (29)

FIG. 21 is a block diagram illustrating an embodiment of a simplifiedadaptive DSP. In this example, DSP 2100 includes a nonlinear adaptivephase filter 2102 and a nonlinear adaptive amplitude filter 2104. Bothfilters receive the same input, and the outputs of the filters arecombined using a combiner 2106. The nonlinear adaptive phase filter isan adaptive filter whose amplitude response is approximately constantover time, but whose phase response varies over time. The nonlinearadaptive amplitude filter is an adaptive filter whose amplitude responseis time-varying, but whose phase response stays approximately constantover time. Thus, the resulting combined filter 2100 has both atime-varying phase response and a time-varying amplitude response. Adesired filter transfer function can be achieved by adjusting thetransfer function of filter 2102, 2104, or both.

FIG. 22 is a block diagram illustrating an embodiment of a nonlinearadaptive phase filter. In this example, nonlinear phase filter 2200 isan infinite impulse response (IIR) filter. As will be described in moredetail below, coefficients ã₀, ã₁, . . . , ã_(k) are nonlinear functionsof appropriate feed-forward and feedback signals in the filter. In theexample shown, only ã₀ and ã₁ are used. The filter structure producesonly phase effects, and the filter output r_(n) has a phase that variesas a nonlinear function of the signal and its history.

An IIR filter has a flat amplitude response and a non-uniform phaseresponse if its z-domain transfer function takes the form

$\begin{matrix}{{H(z)} = \frac{a_{0} + {a_{1}z^{- 1}} + {a_{2}z^{- 2}}}{a_{2} + {a_{1}z^{- 1}} + {a_{0}z^{- 2}}}} & (30)\end{matrix}$or more simply

$\begin{matrix}{{H(z)} = {\frac{a_{0} + {a_{1}z^{- 1}} + z^{- 2}}{1 + {a_{1}z^{- 1}} + {a_{0}z^{- 2}}}.}} & (31)\end{matrix}$The time domain function of such an IIR filter is

$\begin{matrix}\begin{matrix}{r_{n} = {{a_{0}y_{n}} + {a_{1}y_{n - 1}} + y_{n - 2} - {a_{1}r_{n - 1}} - {a_{0}r_{n - 2}}}} \\{= {y_{n - 2} + {a_{0}\left( {y_{n} - r_{n - 2}} \right)} + {{a_{1}\left( {y_{n - 1} - r_{n - 1}} \right)}.}}}\end{matrix} & (32)\end{matrix}$

To achieve the desired nonlinear phase effect, IIR filter 2200 isconfigured to operate according to the following time domain function:r _(n) =y _(n−2) +ã ₀(y _(n) −r _(n−2))+ã ₁(y _(n−1) −r _(n−1))  (33)where each coefficient at is ã_(k) nonlinear function of the inputsignal.

One way of implementing the nonlinear coefficients ã_(k) is

$\begin{matrix}\begin{matrix}{{\overset{\sim}{a}}_{k,n} = {{a_{k,n}s_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{c_{j,n}^{k}{{{{A_{j,n}^{k}}^{T}S_{n}} + \beta_{j,n}^{k}}}}}}} \\{= {{a_{k,n}S_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{{c_{j,n}^{k}\left( {{{A_{j,n}^{k}}^{T}S_{n}} + \beta_{j,n}^{k}} \right)}\lambda_{j,n}^{k}}}}}\end{matrix} & (34)\end{matrix}$where λ_(j,n) ^(k)=sign(A_(j,n) ^(k T)S_(n)+β_(j,n) ^(k)) andS_(n)=[(y_(n)−r_(n−L)(y) _(n−1)−r_(n+L+1)) . . . ].

Other forms can be used, and any nonlinear function will result in anonlinear phase filter with infinite time memory because the filter isan IIR filter. Coefficients of the nonlinear filter are updated asfollows:A _(j,n+1) ^(k T) =A _(j,n) ^(k T) +μc _(j,n) ^(k)λ_(j,n) ^(k) S _(n) e_(n) s _(n)  (35)β_(j,n+1) ^(k)=β_(j,n) ^(k) +μc _(j,n) ^(k)λ_(j,n) ^(k) e _(n) s_(n)  (36)c _(j,n+1) ^(k) =c _(j,n) ^(k) μ|A _(j,n) ^(k T) S _(n)+β_(j,n) ^(k) |e_(n) s  (37)a _(k,n+1) =a _(k,n) +μx _(n) e _(n) s _(n)  (38)b _(k,n+1) =b _(k,n) +μe _(n) s _(n)  (39)

The above example shows an order-2 filter. In some embodiments, filtersof order N are implemented based on the same principle. Such a filterhas a generalized time domain form of:r _(n) =ã ₀(y _(n) −r _(n−N))+ã ₁(y _(n−1) −r _(n−N+1))+ . . . +y_(n−N)  (40)

In some embodiments, nonlinear adaptive amplitude filter 2104 of FIG. 21is implemented using a finite impulse response (FIR) filter to provide atime-varying amplitude response and a phase response that isapproximately constant over time. The FIR filter may be implementedusing any suitable techniques, such as the low complexity filterimplementation techniques described in U.S. patent application Ser. No.11/061,850 by Batruni entitled “LOW COMPLEXITY NONLINEAR FILTERS”, whichis incorporated by reference for all purposes.

In some embodiments, a simplified form of the FIR filter is expressedas:

$\begin{matrix}{u_{n} = {\sum\limits_{i = 0}^{M}{{\overset{\sim}{w}}_{i}y_{n - i}}}} & (41)\end{matrix}$wherein the coefficients {tilde over (w)}_(i) are time-varying functionsof the input signal. The general expression is of order M. For purposesof illustration, examples of order 2 FIRs that depend on threeconsecutive input samples are discussed below.

In some embodiments, the coefficients of a simplified filter aredescribed as follows:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{w}}_{k,n} = {{a_{k,n}y_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{c_{j,n}^{k}{{{{A_{j,n}^{k}}^{T}Y_{n}} + \beta_{j,n}^{k}}}}}}} \\{= {{a_{k,n}y_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{{c_{j,n}^{k}\left( {{{A_{j,n}^{k}}^{T}Y_{n}} + \beta_{j,n}^{k}} \right)}\lambda_{j,n}^{k}}}}}\end{matrix} & (42)\end{matrix}$where K is the number of sub-filters in the nonlinear filter.λ_(j,n) ^(k)=sign(A _(j,n) ^(k T) Y _(n)+β_(j,n) ^(k))  (43)Y_(n)[y_(n+M)y_(n+M−1) . . . y_(n) . . . y_(n−M+1)y_(n−M)]  (44)A_(j,n) ^(k T)=└α_(j,M,n) ^(k)α_(j,M−1,n) ^(k) . . . α_(j,0,n) ^(k) . .. α_(j,−M+1,n) ^(k)α_(j,−M,n) ^(k)┘  (45)The coefficients shown in this example have a time index n because theyare updated over time and therefore are time-varying. Thus, theresulting adaptive filter has a time-varying nonlinear transferfunction. The initial starting values for the coefficients can berandom. In some embodiments small starting values are used to allow thefilter to converge without introducing a great deal of noise during theprocess.

The coefficients of the simplified filter are updated as follows:A _(j,n+1) ^(k T) =A _(j,n) ^(k T) +μc _(j,n) ^(k)λ_(j,n) ^(k) Y _(n) e_(n) y _(n)  (46)β_(j,n+1) ^(k)=β_(j,n) ^(k) +μc _(j,n) ^(k)λ_(j,n) ^(k) e _(n) y_(n)  (47)c _(j,n+1) ^(k) =c _(j,n) ^(k) μ|A _(j,n) ^(k T) Y _(n)+β_(j,n)^(k)|  (48)a _(k,n+1) =a _(k,n) +μy _(n) e _(n) y _(n)  (49)b _(k,n+1) =b _(k,n) +μe _(n) y _(n)  (50)

In some embodiments, the coefficients of a simplified filter aredescribed as follows:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{w}}_{k,n} = {{a_{k,n}y_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{c_{j,n}^{k}\mspace{11mu}\text{sign}\mspace{11mu}\left( {{{A_{j,n}^{k}}^{T}Y_{n}} + \beta_{j,n}^{k}} \right)}}}} \\{= {{a_{k,n}y_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{c_{j,n}^{k}\lambda_{j,n}^{k}}}}}\end{matrix} & (51)\end{matrix}$whereλ_(j,n) ^(k)=sign(A _(j,n) ^(k T) Y _(n)+β_(j,n) ^(k))  (52)Y_(n)[y_(n+M)y_(n+M−1) . . . y_(n) . . . y_(n−M+1)y_(n−M)]  (53)A_(j,n) ^(k T)=[α_(j,M,n) ^(k)α_(j,M−1,n) ^(k) . . . α_(j,0,n) ^(k) . .. α_(j,−M+1,n) ^(k)α_(j,−M,n) ^(k)]  (54)The coefficients, which have a time index n, are time-varying, as is thefilter transfer function of the resulting adaptive nonlinear filter.

The initial values of the filter coefficients may be chosen as small,random values. The filter coefficients are updated as follows:c _(j,n+1) ^(k) =c _(j,n) ^(k)+μλ_(j,n) ^(k) e _(n) y _(n)  (55)a _(k,n+1) =a _(k,n) +μy _(n) e _(n) y _(n)  (56)b _(k,n+1) =b _(k,n) +μe _(n) y _(n)  (57)

In some embodiments, another type of simplified FIR filter having thefollowing form is used:u _(n) ={tilde over (w)} ₀ +{tilde over (w)} ₁ +{tilde over (w)} ₂  (58)where the coefficients are nonlinear functions of the input signal andits history. The input signal itself, however, is not multiplied by thecoefficients. The coefficients are expressed as:

$\begin{matrix}\begin{matrix}{{\overset{\sim}{w}}_{k,n} = {{a_{k,n}y_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{c_{j,n}^{k}\mspace{11mu}\text{sign}\mspace{11mu}\left( {{{A_{j,n}^{k}}^{T}Y_{n}} + \beta_{j,n}^{k}} \right)}}}} \\{= {{a_{k,n}y_{n}} + b_{k,n} + {\sum\limits_{j = 1}^{K}{c_{j,n}^{k}\lambda_{j,n}^{k}}}}}\end{matrix} & (59)\end{matrix}$whereλ_(j,n) ^(k)=sign(A _(j,n) ^(k T) Y _(n)β_(j,n) ^(k))  (60)Y_(n)[y_(n+M)y_(n+M−1) . . . y_(n) . . . y_(n−M+1)y_(n−M)]  (61)A_(j,n) ^(k T)=└α_(j,M,n) ^(k)α_(j,M,n) ^(k) . . . α_(j,0,n) ^(k) . . .α_(j,−M+1,n) ^(k)α_(j,−M,n) ^(k)┘  (62)

The coefficients are updated as follows:c _(j,n+1) ^(k) =c _(j,n) ^(k)+μλ_(j,n) ^(k) e _(n)  (63)a _(k,n+1) =a _(k,n) +μy _(n) e _(n)  (64)b _(k,n+1) =b _(k,n) +μe _(n)  (65)The coefficients have a time index n because the filter is adaptive andtherefore time-varying. The coefficients' initial values are small,random numbers.

Adaptive self-linearization of an unknown distorted signal has beendescribed. The techniques described are generally applicable tononlinear systems. The methods described may be implemented usingfilters, DSPs, as well as implemented as computer code that operates ongeneral purpose processors such as MATLAB™ programs.

Although the foregoing embodiments have been described in some detailfor purposes of clarity of understanding, the invention is not limitedto the details provided. There are many alternative ways of implementingthe invention. The disclosed embodiments are illustrative and notrestrictive.

1. A digital signal processor (DSP) comprising: an input terminalconfigured to receive an input; an adaptive nonlinear phase filtercoupled to the input terminal, the adaptive nonlinear phase filterhaving a time-varying phase response; and an adaptive nonlinearamplitude filter coupled to the input terminal, the adaptive nonlinearamplitude filter having a time-varying amplitude response.
 2. The DSP ofclaim 1, further comprising a combiner to combine the output of theadaptive nonlinear phase filter and the output of the adaptive nonlinearamplitude filter.
 3. The DSP of claim 1, wherein the adaptive nonlinearphase filter has an amplitude response that is approximately constantover time.
 4. The DSP of claim 1, wherein the adaptive nonlinearamplitude filter has a phase response that is approximately constantover time.
 5. The DSP of claim 1, wherein the adaptive nonlinear phasefilter includes an infinite impulse response (IIR) filter.
 6. The DSP ofclaim 1, wherein the adaptive nonlinear phase filter includes aninfinite impulse response (IIR) filter whose time domain function isr_(n)ã₀(y_(n)−r_(n−N))+ã₁(y_(n−1)−r_(n−N+1))+ . . . +y_(n−N).
 7. The DSPof claim 1, wherein the adaptive nonlinear amplitude filter includes afinite impulse response (FIR) filter.
 8. The DSP of claim 7, wherein theFIR filter has a filter coefficient that is a time-varying function ofthe input to the FIR filter.
 9. The DSP of claim 7, wherein the FIRfilter has the form${u_{n} = {\sum\limits_{i = 0}^{M}{{\overset{\sim}{w}}_{i}y_{n - i}}}},$wherein {tilde over (w)} is a time-varying function of the input signal.10. The DSP of claim 8, wherein the time-varying function is a firstorder function of the input to the FIR filter.
 11. A method forprocessing a signal, comprising: receiving the signal as an input to adigital signal processor (DSP); sending the signal to an adaptivenonlinear phase filter, the adaptive nonlinear phase filter having atime-varying phase response; and sending the signal to an adaptivenonlinear amplitude filter, the adaptive nonlinear amplitude filterhaving a time-varying amplitude response.
 12. The method of claim 11,further comprising combining the output of the adaptive nonlinear phasefilter and the output of the adaptive nonlinear amplitude filter. 13.The method of claim 11, wherein the adaptive nonlinear phase filter hasan amplitude, response that is approximately constant over time.
 14. Themethod of claim 11, wherein the adaptive nonlinear amplitude filter hasa phase response that is approximately constant over time.
 15. Themethod of claim 11, wherein the adaptive nonlinear phase filter includesan infinite impulse response (IIR) filter.
 16. The method of claim 11,wherein the adaptive nonlinear phase filter includes an infinite impulseresponse (IIR) filter whose time domain function isr_(n)ã₀(y_(n)−r_(n−N))+ã₁(y_(n−1)−r_(n−N+1))+ . . . +y_(n−N).
 17. Themethod of claim 11, wherein the adaptive nonlinear amplitude filterincludes a finite impulse response (FIR) filter.
 18. The method of claim17, wherein the FIR filter has a filter coefficient that is atime-varying function of the input to the FIR filter.
 19. The method ofclaim 17, wherein the FIR filter has the form${u_{n} = {\sum\limits_{i = 0}^{M}{{\overset{\sim}{w}}_{i}y_{n - i}}}},$wherein {tilde over (w)}_(i) is a time-varying function of the inputsignal.
 20. The method of claim 18, wherein the time-varying function isa first order function of the input to the FIR filter.
 21. A computerprogram product for processing a signal, the computer program productbeing embodied in a computer readable medium and comprising computerinstructions for: receiving the signal; sending the signal to anadaptive nonlinear phase filter, the adaptive nonlinear phase filterhaving a time-varying phase response; and sending the signal to anadaptive nonlinear amplitude filter, the adaptive nonlinear amplitudefilter having a time-varying amplitude response.